Conventional Darlington amplifiers are noted for having a wide band gain and third order intercept point (IP3) response, compact size, and low cost package implementation. Darlington amplifiers typically have superior IP3-bandwidth performance compared to common-source and common-emitter feedback amplifiers. While Darlington amplifiers have advantages over single transistor feedback amplifiers, Darlington amplifiers typically suffer in output power capability due the limited output voltage swing. In particular, the lower voltage swing limit is lower by 1-Vbe, or 1 turn-on voltage, compared to a common-source or common-emitter amplifier. This extra threshold drop is due to the first stage transistor and increases the knee of a typical transistor I-V characteristic by 1 threshold turn-on voltage. As a result, the output P1dB and IP3 will be limited due to the extra threshold drop of the first stage transistor.
Referring to FIG. 1, a diagram of a conventional Darlington feedback amplifier topology 10 is shown. The topology 10 includes the usual Darlington transistor pair 12 (transistor Q1 and transistor Q2) a series feedback resistor RE2, a parallel feedback resistor RFB, and a bias resistor RE1 and a bias resistor RBB. The Darlington pair (or cell) 12 is known to have advantages over the common-emitter or common-source transistor in feedback amplifiers applications. The Darlington cell 12 can provide higher current gain and cut-off frequency, and can be designed to have a higher input impedance, which is preferred in feedback implementations.
The use of the Darlington pair 12 in feedback amplifiers results in improved IP3, Gain and bandwidth performance and can also provide higher output power capability. The drawback of the conventional Darlington topology 10 is that the knee voltage of the I-V characteristics is increased by 1 Vbe, or 1 turn-on threshold voltage, due to the presence of the transistor Q1. In a power amplifier application, the transistor Q1 of the amplifier will saturate 1 Vbe earlier than a common-emitter amplifier.
Referring to FIG. 2, a graph illustrating the difference in I-V characteristics (or current-voltage DC characteristics) of a conventional GaAs HBT Darlington with a GaAs HBT common-emitter device is shown. A common-emitter device illustrates a family of I-V curves that has a Knee voltage of around 0.3V. The Darlington configuration has a knee voltage of ˜2V. The knee voltage of the Darlington is slightly greater than 1 Vbe˜1.45V higher than the knee voltage of the common-emitter device as theory would predict. Therefore, the output voltage swing of the Darlington cannot swing below ˜2V without saturating the first transistor Q1. Such a swing limitation inhibits the linear output power capability of a Darlington based amplifier and will impact linearity figure of merits such as IP3 and adjacent channel power rejection (ACPR) (i.e., both distortion measurements).
Referring to FIG. 3, a system 20 illustrates a modification of the Darlington feedback amplifier. An example of such a configuration may be found in (i) the Agilent HMMC-5200 20 GHz HBT Series-Shunt Amplifier data sheet, and (ii) the RF Nitro Communications Inc. NDA-212 GaInP/GaAs HBT MMIC Distributed Amplifier DC-17 GHz. The first stage collector is not connected to the output, but instead is connected to a separate supply through an optional stability resistor. The power supply is conventionally bypassed with a capacitor. The knee voltage of the I-V curves of this Pseudo-Darlington is now reduced to that of the common-emitter output transistor Q2. Such a topology is not a true Darlington because the collectors of the input and output transistors are not tied together. Therefore, the RF and functional characteristics of such a CC-CE amplifier will depart from the true Darlington feedback amplifier.
Although the knee voltage of the CC-CE amplifier is lower and should obtain better output power, such an amplifier uses an additional supply voltage. Such an additional supply voltage precludes use in popular transistor style packages which have only an input, output and ground lead, such as the SOT89, SOT86 and 76 ceramic pill packages. Additionally, the stability nature of the CC-CE topology is marginal due to the common-collector input stage. The instability of the common-collector is well known in analog design disciplines. The common collector results in a positive gain slope, which is attributed to modest positive feedback inherent to the topology. This results in positive gain slope, marginal stability, and degraded IP3 and NF performance as compared to the Darlington amplifier of FIG. 1. What is achieved by using this CC-CE topology is a larger small signal gain-bandwidth product, which is useful for small signal, low power applications. One such application that has popularized this topology is the transimpedance amplifier for fiber optic receiver applications, which uses a wide small signal bandwidth response. Although practical for small signal applications, such a system is less than optimal for linear output power applications.
It would be desirable to implement a method and/or apparatus for reducing the knee voltage and improving the output power capability of a Darlington by implementing a combination of E-mode and D-mode PHEMT devices.